1. Field of the Invention
The present invention relates to a current-mode controlled switching regulator and a control method therefor.
2. Description of the Related Art
Until recently, a switching regulator has generally employed a voltage-mode control system, in which a switching device is controlled by PWM (pulse width modulation) control based on the voltage difference between an output voltage and a reference voltage to stabilize the output voltage. However, since such a voltage-mode controlled switching regulator detects a feedback signal from the output voltage, response to fluctuation in the output voltage is slow. Further, phase compensation for an error amplifier that amplifies the voltage difference between the output voltage and the reference voltage becomes complicated.
To overcome these drawbacks, a current-mode controlled switching regulator is increasingly used. However, it is known that subharmonic oscillation may occur using the current-mode controlled switching regulator and the current-mode controlled switching regulator may become uncontrollable when a pulse width modulated signal has a duty cycle of more than 50%. The subharmonic oscillation is generally avoided by adding slope compensation to the PWM control.
FIG. 1 is a diagram illustrating example circuitry of a current-mode controlled switching regulator with a slope compensation circuit, which is a step-down switching regulator having an input terminal IN to which an input voltage Vin is applied and an output terminal OUT from which an output voltage Vout is output.
In FIG. 1, when a switching transistor 105 is turned on, power is supplied to an inductor 104, a smoothing capacitor 102, and a load 101. When the switching transistor 105 is turned off, the energy stored in the inductor 104 and the smoothing capacitor 102 is supplied to the load 101. A current-to-voltage converter 106 has an impedance Rsense for converting a current iL flowing through the inductor 104 into a converted voltage Vsense, or Rsense×iL.
An oscillator 110 generates and outputs a reference clock signal CLK and a sawtooth voltage signal Vramp. An adder 108 adds the sawtooth voltage signal Vramp to the converted voltage Vsense for slope compensation and outputs a slope voltage Vs to a non-inverted input terminal of a PWM comparator 107. An error amplifier 115 amplifies the voltage difference between a divided voltage Vfb generated by dividing the output voltage Vout and a reference voltage Vref and generates and outputs an error voltage Ve to an inverted input terminal of the PWM comparator 107. The PWM comparator 107 compares the error voltage Ve with the slope voltage Vs. When the error voltage Ve exceeds the slope voltage Vs, the PWM comparator 107 resets an RS latch circuit 112 and turns off the switching transistor 105. Accordingly, the peak current value of the inductor current iL depends on the error voltage Ve.
The output voltage Vout is controlled as follows. When the divided voltage Vfb is higher than the reference voltage Vref, the error voltage Ve is lowered to lower the output voltage Vout. When the divided voltage Vfb is lower than the reference voltage Vref, the error voltage Ve is increased to increase the output voltage Vout.
To avoid subharmonic oscillation by slope compensation, it is desirable that the slope angle of the slope voltage Vs be not less than half the slope angle of the inductor current iL flowing when the switching transistor 105 is off.
Specifically, the slope diL/dt of the inductor current iL flowing in the current-mode controlled switching regulator of FIG. 1 is expressed by the following formula (a) when the switching transistor 105 is on, and expressed by the following formula (b) when the switching transistor 105 is off:diL/dt=(Vin−Vout)/L  (a)diL/dt=−Vout/L  (b),
where L is the inductance of the inductor 104.
The slope angle of the sawtooth voltage signal Vramp, which is referred to as a slope compensation value Iramp, is expressed by the following formula (c):Iramp>Vout/2/L×Rsense  (c)
The following formulas (d), (e), and (f) can be provided for a step-up switching regulator, corresponding to the formulas (a), (b), and (c), respectively.diL/dt=Vin/L  (d)diL/dt=−(Vout−Vin)/L  (e)Iramp>(Vout−Vin)/L/2×Rsense  (f)
The slope compensation value Iramp is expressed by using the input voltage Vin and the output voltage Vout without problem when the input voltage Vin and the output voltage Vout are fixed. However, the input voltage Vin and the output voltage Vout generally fluctuate in a wide range. When the slope compensation value Iramp is fixed, it is necessary that the slope compensation value Iramp is set to be a value that is a maximum value in the estimated fluctuation range of the input voltage Vin and the output voltage Vout. By performing excessive slope compensation like this case, subharmonic oscillation can be avoided. However, the effect of current feedback decreases and the operation becomes similar to the operation of the voltage-mode control system. Consequently, controllability decreases. Therefore, to perform adequate slope compensation in a wide input/output voltage range, the amount of slope compensation is determined based on input and output voltage levels.
However, in this case, since the amount of slope compensation is changed according to the input and output voltages, the circuitry is complicated. In addition, a general-purpose IC for a switching regulator generally uses an external resistor to generate a divided voltage by dividing the output voltage, which may prevent monitoring of the output voltage and thereby prevent slope compensation in accordance with the output voltage.